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NCP1216, NCP1216A PWM Current-Mode Controller for High-Power Universal Off-Line Supplies
Housed in a SOIC-8 or PDIP-7 package, the NCP1216 represents an enhanced version of NCP1200 based controllers. Due to its high drive capability, NCP1216 drives large gate-charge MOSFETs, which together with internal ramp compensation and built-in frequency jittering, ease the design of modern AC-DC adapters. With an internal structure operating at different fixed frequencies, the controller supplies itself from the high-voltage rail, avoiding the need of an auxiliary winding. This feature naturally eases the designer task in some particular applications, e.g. battery chargers or TV sets. Current-mode control also provides an excellent input audio susceptibility and inherent pulse-by-pulse control. Internal ramp compensation easily prevents sub-harmonic oscillations from taking place in continuous conduction mode designs. When the current setpoint falls below a given value, e.g. the output power demand diminishes, the IC automatically enters the so-called skip cycle mode and provides excellent efficiency at light loads. Because this occurs at a user adjustable low peak current, no acoustic noise takes place. The NCP1216 features an efficient protective circuitry, which in presence of an over current condition disables the output pulses while the device enters a safe burst mode, trying to restart. Once the default has gone, the device auto-recovers.
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MINIATURE PWM CONTROLLER FOR HIGH-POWER AC-DC WALL ADAPTERS AND OFF-LINE BATTERY CHARGERS
MARKING DIAGRAMS
8 SOIC-8 D SUFFIX CASE 751 1 16XXX ALYW
PDIP-7 P SUFFIX CASE 626B 1 XXXX A WL, L YY, Y WW, W = = = = =
P1216XXXX AWL YYWW
* * * * * * * * * * * * * * * * * * *
Pb-Free Package is Available No Auxiliary Winding Operation Current-Mode Control with Adjustable Skip-Cycle Capability Internal Ramp Compensation Limited Duty Cycle to 50% (A version only) Internal 1.0 ms Soft-Start (A version only) Built-In Frequency Jittering for Better EMI Signature Auto-Recovery Internal Output Short-Circuit Protection Extremely Low No-Load Standby Power 500 mA Peak Current Capability Fixed Frequency Versions at 65 kHz, 100 kHz, 133 kHz Internal Temperature Shutdown Direct Optocoupler Connection SPICE Models Available for TRANsient and AC Analysis Pin-to-Pin Compatible with NCP1200 Series
Specific Device Code Assembly Location Wafer Lot Year Work Week
DEVICE MARKING INFORMATION
See detailed device marking information in the ordering information section on page 16 of this data sheet.
PIN CONNECTIONS
Adj 1 FB 2 CS 3 Gnd 4 8 HV 7 NC 6 VCC 5 Drv
Typical Applications
High Power AC-DC Converters for TVs, Set-Top Boxes, etc. Offline Adapters for Notebooks Telecom DC-DC Converters All Power Supplies
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ORDERING INFORMATION
See detailed ordering and shipping information in the ordering information section on page 16 of this data sheet.
(c) Semiconductor Components Industries, LLC, 2004
September, 2004 - Rev. 8
Publication Order Number: NCP1216/D
NCP1216, NCP1216A
+ *See Application Section NCP1216 + EMI Filter
= 35kHz 1 Fosc Adj HV 8 2 FB 7
3 4
CS Vcc 6 GNDDrv 5 Rcomp
Universal Input + Rsense
Figure 1. Typical Application Example
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PIN FUNCTION DESCRIPTION
Pin No. 1 Pin Name Adj Function Pin Description Adjust the Skipping Peak Current This pin lets you adjust the level at which the cycle skipping process takes place. Shorting this pin to ground, permanently disables the skip cycle feature. By connecting an Optocoupler to this pin, the peak current setpoint is adjusted accordingly to the output power demand. This pin senses the primary current and routes it to the internal comparator via an L.E.B. By inserting a resistor in series with the pin, you control the amount of ramp compensation you need. - The driver's output to an external MOSFET. This pin is connected to an external bulk capacitor of typically 22 mF. This un-connected pin ensures adequate creepage distance. Connected to the high-voltage rail, this pin injects a constant current into the VCC bulk capacitor. 2 3 FB CS Sets the Peak Current Setpoint Current Sense Input 4 5 6 7 8 GND Drv VCC NC HV IC Ground Driving Pulses Supplies the IC - Generates the VCC from the Line
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NCP1216, NCP1216A
Adj
1
HV Current Source 96 k 1.1 V Skip Cycle Comparator Internal VCC + - Clock Jittering
8 HV
FB
2
UVLO High and Low Internal Regulator
7 NC
25 k
Current 3 Sense
220 ns L.E.B 19 k 20 k Ramp Compensation
65 kHz 100 kHz 133kHz + - 1V
Set Q Flip-Flop Q DCmax = 75% Reset $500 mA 1 ms SS*
6 VCC
GND
4
5 Drv
Pull-up Resistor 57 k + Vref 25 k - 5V
Overload? Fault Duration
* Available for "A" version only.
Figure 2. Internal Circuit Architecture
MAXIMUM RATINGS
Rating Power Supply Voltage, VCC Pin Maximum Voltage on Low Power Pins (except Pin 8 and Pin 6) Maximum Voltage on Pin 8 (HV), Pin 6 (VCC) Decoupled to Ground with 10 mF Maximum Voltage on Pin 8 (HV), Pin 6 (VCC) Grounded Minimum Operating Voltage on Pin 8 (HV) Maximum Current into all Pins except VCC (Pin 6) and HV (Pin 8) when 10 V ESD Diodes are Activated Thermal Resistance Junction-to-Air, PDIP-7 Version Thermal Resistance Junction-to-Air, SOIC-8 Version Maximum Junction Temperature Temperature Shutdown Hysteresis in Shutdown Storage Temperature Range ESD Capability, HBM Model (All Pins except VCC and HV) ESD Capability, Machine Model RqJ-A RqJ-A TJMAX TSD Symbol VCC Value 16 -0.3 to 10 500 450 30 5.0 100 178 150 155 30 -60 to +150 2.0 200 Unit V V V V V mA C/W C C C C kV V
Maximum ratings are those values beyond which device damage can occur. Maximum ratings applied to the device are individual stress limit values (not normal operating conditions) and are not valid simultaneously. If these limits are exceeded, device functional operation is not implied, damage may occur and reliability may be affected.
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ELECTRICAL CHARACTERISTICS (For typical values TJ = 25C, for min/max values TJ = 0C to +125C, Maximum TJ = 150C,
VCC = 11 V unless otherwise noted.) Characteristic DYNAMIC SELF-SUPPLY VCC Increasing Level at which the Current Source Turns Off VCC Decreasing Level at which the Current Source Turns On VCC Decreasing Level at which the Latchoff Phase Ends Internal IC Consumption, No Output Load on Pin 5, FSW = 65 kHz Internal IC Consumption, No Output Load on Pin 5, FSW = 100 kHz Internal IC Consumption, No Output Load on Pin 5, FSW = 133 kHz Internal IC Consumption, 1.0 nF Output Load on Pin 5, FSW = 65 kHz Internal IC Consumption, 1.0 nF Output Load on Pin 5, FSW = 100 kHz Internal IC Consumption, 1.0 nF Output Load on Pin 5, FSW = 133 kHz Internal IC Consumption, Latchoff Phase, VCC = 6.0 V INTERNAL STARTUP CURRENT SOURCE (TJ > 0C) High-voltage Current Source, VCC = 10 V High-voltage Current Source, VCC = 0 V DRIVE OUTPUT Output Voltage Rise-time @ CL = 1.0 nF, 10-90% of a 12 V Output Signal Output Voltage Fall-time @ CL = 1.0 nF, 10-90% of a 12 V Output Signal Source Resistance Sink Resistance CURRENT COMPARATOR (Pin 5 Unloaded) Input Bias Current @ 1.0 V Input Level on Pin 3 Maximum Internal Current Setpoint Default Internal Current Setpoint for Skip Cycle Operation Propagation Delay from Current Detection to Gate OFF State Leading Edge Blanking Duration 1. VCCOFF and VCCON min-max always ensure an hysteresis of 2.0 V. 2. Maximum value at TJ = 0C. 3. Minimum value for TJ = 125C. 3 3 3 3 3 IIB ILimit ILskip TDEL TLEB 0.93 0.02 1.08 330 80 220 130 1.14 mA V mV ns ns 5 5 5 5 Tr Tf ROH ROL 15 5.0 60 20 20 10 35 18 ns ns W W 8 8 IC1 IC2 4.9 (Note 3) 8.0 9.0 11 mA mA NCP1216 NCP1216A 6 6 6 6 6 6 6 6 6 6 VCCOFF VCCON VCClatch ICC1 ICC1 ICC1 ICC2 ICC2 ICC2 ICC3 11.2 9.2 12.2 10.0 5.6 990 1025 1060 1.7 2.1 2.4 250 320 1110 (Note 2) 1180 (Note 2) 1200 (Note 2) 2.0 (Note 2) 2.4 (Note 2) 2.9 (Note 2) 13.4 (Note 1) 11.0 (Note 1) V V V mA mA mA mA mA mA mA Pin Symbol Min Typ Max Unit
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ELECTRICAL CHARACTERISTICS (continued) (For typical values TJ = 25C, for min/max values TJ = 0C to +125C, Maximum
TJ = 150C, VCC = 11 V unless otherwise noted.) Characteristic INTERNAL OSCILLATOR (VCC = 11 V, Pin 5 Loaded by 1.0 kW) Oscillation Frequency, 65 kHz Version Oscillation Frequency, 100 kHz Version Oscillation Frequency, 133 kHz Version Built-in Frequency Jittering in Percentage of fOSC Maximum Duty-Cycle FEEDBACK SECTION (VCC = 11 V, Pin 5 Loaded by 1.0 kW) Internal Pullup Resistor Pin 2 (FB) to Internal Current Setpoint Division Ratio SKIP CYCLE GENERATION Default Skip Mode Level Pin 1 Internal Output Impedance INTERNAL RAMP COMPENSATION Internal Ramp Level @ 25C (Note 4) Internal Ramp Resistance to CS Pin 4. A 1.0 MW resistor is connected to the ground for the measurement. 3 3 Vramp Rramp 2.6 2.9 19 3.2 V kW 1 1 Vskip Zout 0.9 1.1 25 1.26 V kW 2 - Rup Iratio 20 3.3 kW NCP1216 NCP1216A fOSC fOSC fOSC fjitter Dmax 69 42 58.5 90 120 65 100 133 4.0 75 46.5 81 50 71.5 110 146 kHz kHz kHz % % Pin Symbol Min Typ Max Unit
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TYPICAL CHARACTERISTICS
HV PIN LEAKAGE CURRENT @ 500 V (mA) 50 14.0 13.5 13.0 12.5 12.0 11.5 11.0 -25
40 VCCOFF (V)
30
20
10
0 -25
0
25
50
75
100
125
0
25
50
75
100
125
TEMPERATURE (C)
TEMPERATURE (C)
Figure 3. High Voltage Pin Leakage Current vs. Temperature
12.0 11.5 11.0 VCCON (V) ICC1 (mA) 10.5 10.0 9.5 9.0 -25 1400 1300 1200 1100 1000 900 800 700 600 0 25 50 75 100 125 500 -25
Figure 4. VCCOFF vs. Temperature
133 kHz
65 kHz
100 kHz
0
25
50
75
100
125
TEMPERATURE (C)
TEMPERATURE (C)
Figure 5. VCCON vs. Temperature
Figure 6. ICC1 (@ VCC = 11 V) vs. Temperature
2.80 2.60 2.40 2.20 ICC2 (mA) 2.00 1.80 1.60 1.40 1.20 1.00 -25 0 25 50 75 TEMPERATURE (C) 100 125 65 kHz 133 kHz FOSC (kHz) 100 kHz
150
130
133 kHz
110 100 kHz 90
70
65 kHz
50 -25
0
25 50 75 TEMPERATURE (C)
100
125
Figure 7. ICC2 vs. Temperature
Figure 8. Switching Frequency vs. Temperature
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5.90 5.80 5.70 ICC3 (mA) 250 200 150 100 5.40 50 5.30 -25 0 25 50 75 100 125 0 -25 0 25 50 75 100 125 NCP1216 5.60 5.50 400 350 300 VCClatch (V) NCP1216A
TEMPERATURE (C)
TEMPERATURE (C)
Figure 9. VCClatch vs. Temperature
Figure 10. ICC3 vs. Temperature
30 DRIVER RESISTANCE (W) 25 Source 20 15 10 5 0 -25 Sink CURRENT SENSE LIMIT (V)
1.13
1.08
1.03
0.98
0
25
50
75
100
125
0.93 -25
0
25
50
75
100
125
TEMPERATURE (C)
TEMPERATURE (C)
Figure 11. Drive Sink and Source Resistance vs. Temperature
Figure 12. Current Sense Limit vs. Temperature
1.20
75.0 74.5 DUTY CYCLE (%) 0 25 50 75 100 125
1.15 Vskip (V)
74.0 73.5 73.0 72.5
1.10
1.05
1.00 -25
72.0 -25
0
25
50
75
100
125
TEMPERATURE (C)
TEMPERATURE (C)
Figure 13. Vskip vs. Temperature
Figure 14. NCP1216 Max Duty-Cycle vs. Temperature http://onsemi.com
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49.0 48.5 DUTY CYCLE (%) 48.0 Vramp (V) 0 25 50 75 100 125 47.5 47.0 46.5 46.0 45.5 45.0 -25 3.10 3.05 3.00 2.95 2.90 2.85 2.80 2.75 2.70 -25
0
25
50
75
100
125
TEMPERATURE (C)
TEMPERATURE (C)
Figure 15. NCP1216A Max Duty-Cycle vs. Temperature
Figure 16. Vramp vs. Temperature
14 12 10 IC1 (mA) 8 6 4 2 -25
0
25
50
75
100
125
TEMPERATURE (C)
Figure 17. High Voltage Current Source (@ VCC = 10 V) vs. Temperature
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APPLICATION INFORMATION
Introduction
The NCP1216 implements a standard current mode architecture where the switch-off event is dictated by the peak current setpoint. This component represents the ideal candidate where low part count is the key parameter, particularly in low-cost AC-DC adapters, TV power supplies etc. Due to its high-performance High-Voltage technology, the NCP1216 incorporates all the necessary components normally needed in UC384X based supplies: timing components, feedback devices, low-pass filter and self-supply. This later point emphasizes the fact that ON Semiconductor's NCP1216 does NOT need an auxiliary winding to operate: the product is naturally supplied from the high-voltage rail and delivers a VCC to the IC. This system is called the Dynamic Self-Supply (DSS): Dynamic Self-Supply (DSS): Due to its Very High Voltage Integrated Circuit (VHVIC) technology, ON Semiconductor's NCP1216 allows for a direct pin connection to the high-voltage DC rail. A dynamic current source charges up a capacitor and thus provides a fully independent VCC level to the NCP1216. As a result, there is no need for an auxiliary winding whose management is always a problem in variable output voltage designs (e.g. battery chargers). Adjustable Skip Cycle Level: By offering the ability to tailor the level at which the skip cycle takes place, the designer can make sure that the skip operation only occurs at low peak current. This point guarantees a noise-free operation with cheap transformers. Skip cycle offers a proven mean to reduce the standby power in no or light loads situations. Internal Frequency Dithering for Improved EMI Signature: By modulating the internal switching frequency with the DSS VCC ripple, natural energy spread appears and softens the controller's EMI signature. Wide Switching - Frequency Offered with Different Options (65 kHz - 100 kHz - 133 kHz): Depending on the application, the designer can pick up the right device to help reducing magnetics or improve the EMI signature before reaching the 150 kHz starting point. Ramp Compensation: By inserting a resistor between the Current Sense (CS) pin and the actual sense resistor, it becomes possible to inject a given amount of ramp compensation since the internal sawtooth clock is routed to the CS pin. Sub-harmonic oscillations in Continuous Conduction Mode (CCM) can thus be compensated via a single resistor.
Over Current Protection (OCP): By continuously monitoring the FB line activity, NCP1216 enters burst mode as soon as the power supply undergoes an overload. The device enters a safe low power operation, which prevents from any lethal thermal runaway. As soon as the default disappears, the power supply resumes operation. Unlike other controllers, overload detection is performed independently of any auxiliary winding level. In presence of a bad coupling between both power and auxiliary windings, the short circuit detection can be severely affected. The DSS naturally shields you against these troubles. Wide Duty-Cycle Operation: Wide mains operation requires
a large duty-cycle excursion. The NCP1216 can go up to 75% typically. For Continuous Conduction Mode (CCM) applications, the internal ramp compensation lets you fight against sub-harmonic oscillations.
Low Standby Power: If SMPS naturally exhibit a good efficiency at nominal load, they begin to be less efficient when the output power demand diminishes. By skipping unnecessary switching cycles, the NCP1216 drastically reduces the power wasted during light load conditions. In no-load conditions, the NPC1216 allows the total standby power to easily reach next International Energy Agency (IEA) recommendations. No Acoustic Noise While Operating: Instead of skipping cycles at high peak currents, the NCP1216 waits until the peak current demand falls below a user-adjustable 1/3 of the maximum limit. As a result, cycle skipping can take place without having a singing transformer, one can thus select cheap magnetic components free of noise problems. External MOSFET Connection: By leaving the external MOSFET external to the IC, you can select avalanche proof devices, which in certain cases (e.g. low output powers), let you work without an active clamping network. Also, by controlling the MOSFET gate signal flow; you have an option to slow down the device commutation, therefore reducing the amount of ElectroMagnetic Interference (EMI). SPICE Model: A dedicated model to run transient cycle-by-cycle simulations is available but also an averaged version to help you closing the loop. Ready-to-use templates can be downloaded in OrCAD's PSpice and INTUSOFT's IsSpice from ON Semiconductor web site, in the NCP1216 related section.
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Dynamic Self-Supply
The DSS principle is based on the charge/discharge of the VCC bulk capacitor from a low level up to a higher level. We can easily describe the current source operation with a bunch of simple logical equations: POWER-ON: If VCC < VCCOFF then the Current Source is ON, no output pulses If VCC decreasing > VCCON then the Current Source is OFF, output is pulsing If VCC increasing < VCCOFF then the Current Source is ON, output is pulsing Typical values are: VCCOFF = 12.2 V, VCCON = 10 V To better understand the operational principle, Figure 18 offers the necessary light:
Vripple = 2.2 V
Application note AND8069/D details tricks to widen the NCP1216 driving implementation, in particular for large Qg MOSFETs. This document can be downloaded at www.onsemi.com/pub/Collateral/AND8069-D.PDF.
Ramp Compensation
Ramp compensation is a known mean to cure sub-harmonic oscillations. These oscillations take place at half the switching frequency and occur only during Continuous Conduction Mode (CCM) with a duty-cycle greater than 50%. To lower the current loop gain, one usually injects between 50% and 100% of the inductor down-slope. Figure 19 depicts how internally the ramp is generated:
DCmax = 75C 2.9V
VCCOFF = 12.2 V
0V
VCCON = 10 V 19 k ON, I = 8 mA OFF, I = 0 mA L.E.B CS Rsense - + Rcomp
Output Pulse 10 30 50 70 90
From Set-point
Figure 18. The Charge/Discharge Cycle Over a 10 mF VCC Capacitor
Figure 19. Inserting a Resistor in Series with the Current Sense Information brings Ramp Compensation
The DSS behavior actually depends on the internal IC consumption and the MOSFET's gate charge Qg. If we select a 600 V 10 A MOSFET featuring a 30 nC Qg, then we can compute the resulting average consumption supported by the DSS which is:
Itotal [ Fsw Qg ) ICC1.
(eq. 1)
In the NCP1216, the ramp features a swing of 2.9 V with a Duty cycle max at 75%. Over a 65 kHz frequency, it corresponds to a
2.9 0.75 65 kHz + 251 mV ms ramp.
(eq. 5)
The total IC heat dissipation incurred by the DSS only is given by:
Itotal Vpin8.
(eq. 2)
In our FLYBACK design, let's suppose that our primary inductance Lp is 350 mH, delivering 12 V with a Np : Ns ratio of 1:0.1. The OFF time primary current slope is thus given by:
Vout ) Vf Lp Np Ns + 371 mA ms or37 mV ms
(eq. 6)
Suppose that we select the NCP1216P065 with the above MOSFET, the total current is
(30 n 65 k) ) 900 m + 2.9 mA.
(eq. 3)
Supplied from a 350 VDC rail (250 VAC), the heat dissipated by the circuit would then be:
350 V 2.9 mA + 1 W
(eq. 4)
when projected over an Rsense of 0.1 W, for instance. If we select 75% of the down-slope as the required amount of ramp compensation, then we shall inject 27 mV/ms. Our internal compensation being of 251 mV/ms, the divider ratio (divratio) between Rcomp and the 19 kW is 0.107. A few lines of algebra to determine Rcomp:
19 k divratio + 2.37 kW 1 * divratio Frequency Jittering
(eq. 7)
As you can see, it exists a tradeoff where the dissipation capability of the NCP1216 fixes the maximum Qg that the circuit can drive, keeping its dissipation below a given target. Please see the "Power Dissipation" section for a complete design example and discover how a resistor can help to heal the NCP1216 heat equation.
Frequency jittering is a method used to soften the EMI signature by spreading the energy in the vicinity of the main switching component. NCP1216 offers a $4% deviation of
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NCP1216, NCP1216A
the nominal switching frequency whose sweep is synchronized with the VCC ripple. For instance, with a 2.2 V peak-to-peak ripple, the NCP1216P065 frequency will equal 65 kHz in the middle of the ripple and will increase as VCC rises or decrease as VCC ramps down. Figure 20 portrays the behavior we have adopted:
VCCOFF VCC Ripple
To better understand how this skip cycle mode takes place, a look at the operation mode versus the FB level immediately gives the necessary insight:
FB
68 kHz 4.2 V, FB Pin Open
65 kHz
Normal Current Mode Operation Skip Cycle Operation IpMIN = 333 mV / Rsense
3.2 V, Upper Dynamic Range
1V
62 kHz VCCON
Figure 21.
Figure 20. VCC Ripple is Used to Introduce a Frequency Jittering on the Internal Oscillator Sawtooth
Skipping Cycle Mode
The NCP1216 automatically skips switching cycles when the output power demand drops below a given level. This is accomplished by monitoring the FB pin. In normal operation, pin 2 imposes a peak current accordingly to the load value. If the load demand decreases, the internal loop asks for less peak current. When this setpoint reaches a determined level, the IC prevents the current from decreasing further down and starts to blank the output pulses: the IC enters the so-called skip cycle mode, also named controlled burst operation. The power transfer now depends upon the width of the pulse bunches (Figure 22). Suppose we have the following component values: Lp, primary inductance = 350 mH Fsw, switching frequency = 65 kHz Ip skip = 600 mA (or 333 mV / Rsense) The theoretical power transfer is therefore:
1 2 Lp Ip2 Fsw + 4 W.
(eq. 8)
When FB is above the skip cycle threshold (1.0 V by default), the peak current cannot exceed 1.0 V/Rsense. When the IC enters the skip cycle mode, the peak current cannot go below Vpin1 / 3.3. The user still has the flexibility to alter this 1.0 V by either shunting pin 1 to ground through a resistor or raising it through a resistor up to the desired level. Grounding pin 1 permanently invalidates the skip cycle operation.
Power P1
Power P2
Power P3
Figure 22. Output Pulses at Various Power Levels (X = 5 ms/div) P1 < P2 < P3
If this IC enters skip cycle mode with a bunch length of 10 ms over a recurrent period of 100 ms, then the total power transfer is:
4 0.1 + 400 mW.
(eq. 9)
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300 Max Peak Current Skip Cycle Current Limit
200
due to the DSS operation. In our example, at Tambient = 50C, ICC2 is measured to be 2.9 mA with a 10 A / 600 V MOSFET. As a result, the NCP1216 will dissipate from a 250 VAC network,
350 V 2.9 mA@TA + 50C + 1 W
(eq. 11)
100
0
315.4U
882.7U
1.450M
2.017M
2.585M
Figure 23. The Skip Cycle Takes Place at Low Peak Currents which Guarantees Noise Free Operation
The PDIP-7 package offers a junction-to-ambient thermal resistance RqJ-A of 100C/W. Adding some copper area around the PCB footprint will help decreasing this number: 12 mm x 12 mm to drop RqJ-A down to 75C/W with 35 m copper thickness (1 oz.) or 6.5 mm x 6.5 mm with 70 m copper thickness (2 oz.). For a SOIC-8, the original 178C/W will drop to 100C/W with the same amount of copper. With this later PDIP-7 number, we can compute the maximum power dissipation that the package accepts at an ambient of 50C:
T * TAmax P max + Jmax +1W RqJ * A
(eq. 12)
Non-Latching Shutdown
In some cases, it might be desirable to shut off the part temporarily and authorize its restart once the default has disappeared. This option can easily be accomplished through a single NPN bipolar transistor wired between FB and ground. By pulling FB below the Adj pin 1 level, the output pulses are disabled as long as FB is pulled below pin 1. As soon as FB is relaxed, the IC resumes its operation. Figure 24 depicts the application example:
1 2 3 ON/OFF Q1 4 8 7 6 5
which barely matches our previous budget. Several solutions exist to help improving the situation: 1. Insert a Resistor in Series with Pin 8: This resistor will take a part of the heat normally dissipated by the NCP1216. Calculations of this resistor imply that Vpin8 does not drop below 30 V in the lowest mains conditions. Therefore, Rdrop can be selected with:
V * 50 V Rdrop v bulkmin 8 mA
(eq. 13)
In our case, Vbulk minimum is 120 VDC, which leads to a dropping resistor of 8.7 kW. With the above example in mind, the DSS will exhibit a duty-cycle of:
2.9 mA 8 mA + 36%
(eq. 14)
By inserting the 8.7 kW resistor, we drop
8.7 kW * 8 mA + 69.6 V
(eq. 15)
during the DSS activation. The power dissipated by the NCP1216 is therefore:
Figure 24. Another Way of Shutting Down the IC without a Definitive Latchoff State Pinstant * DSSduty * cycle + (350 * 69) * 8 m * 0.36 + 800 mW 1 W * 800 mW + 200 mW
(eq. 16)
A full latching shutdown, including overtemperature protection, is described in application note AND8069/D.
Power Dissipation
We can pass the limit and the resistor will dissipate
(eq. 17)
or
2 pdrop + 69 * 0.36 8.7 k
(eq. 18)
The NCP1216 is directly supplied from the DC rail through the internal DSS circuitry. The current flowing through the DSS is therefore the direct image of the NCP1216 current consumption. The total power dissipation can be evaluated using:
(VHVDC * 11 V) ICC2
(eq. 10)
which is, as we saw, directly related to the MOSFET Qg. If we operate the device on a 90-250 VAC rail, the maximum rectified voltage can go up to 350 VDC. However, as the characterization curves show, the current consumption drops at a higher junction temperature, which quickly occurs
2. Select a MOSFET with a Lower Qg : Certain MOSFETs exhibit different total gate charges depending on the technology they use. Careful selection of this component can help to significantly decrease the dissipated heat. 3. Implement Figure 3, from AN8069/D, Solution: This is another possible option to keep the DSS functionality (good short-circuit protection and EMI jittering) while driving any types of MOSFETs. This solution is recommended when the designer plans to use SOIC-8 controllers.
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4. Connect an Auxiliary Winding: If the mains conditions are such that you simply can't match the maximum power dissipation, then you need to connect an auxiliary winding to permanently disconnect the startup source.
Overload Operation
In applications where the output current is purposely not controlled (e.g. wall adapters delivering raw DC level), it is interesting to implement a true short-circuit protection. A short-circuit actually forces the output voltage to be at a low level, preventing a bias current to circulate in the Optocoupler LED. As a result, the FB pin level is pulled up to 4.2 V, as internally imposed by the IC. The peak current setpoint goes to the maximum and the supply delivers a rather high power with all the associated effects. Please note that this can also happen in case of feedback loss, e.g. a broken Optocoupler. To account for this situation, NCP1216 hosts a dedicated overload detection circuitry. Once activated, this circuitry imposes to deliver pulses in a burst manner with a low duty-cycle. The system auto-recovers when the fault condition disappears.
VCC Regulation Occurs Here 12.2 V 10 V 5.6 V
During the startup phase, the peak current is pushed to the maximum until the output voltage reaches its target and the feedback loop takes over. This period of time depends on normal output load conditions and the maximum peak current allowed by the system. The time-out used by this IC works with the VCC decoupling capacitor: as soon as the VCC decreases from the VCCOFF level (typically 12.2 V) the device internally watches for an overload current situation. If this condition is still present when the VCCON level is reached, the controller stops the driving pulses, prevents the self-supply current source to restart and puts all the circuitry in standby, consuming as little as 350 mA typical (ICC3 parameter). As a result, the VCC level slowly discharges toward 0 V. When this level crosses 5.6 V typical, the controller enters a new startup phase by turning the current source on: VCC rises toward 12.2 V and again delivers output pulses at the VCCOFF crossing point. If the fault condition has been removed before VCCON approaches, then the IC continues its normal operation. Otherwise, a new fault cycle takes place. Figure 25 shows the evolution of the signals in presence of a fault.
Latchoff Phase
Time Drv
VCCOFF = 12.2 V VCCON = 10 V VCClatch = 5.6 V
Driver Pulses Internal Fault Flag
Driver Pulses Time
Fault is Relaxed Time Startup Phase Fault Occurs Here
Figure 25.
If the fault is relaxed during the VCC natural fall down sequence, the IC automatically resumes. If the fault still persists when VCC reached VCCON, then the controller cuts everything off until recovery.
Calculating the VCC Capacitor
As the above section describes, the fall down sequence depends upon the VCC level: how long does it take for the VCC line to go from 12.2 V to 10 V. The required time depends on the startup sequence of your system, i.e. when
you first apply the power to the IC. The corresponding transient fault duration due to the output capacitor charging must be less than the time needed to discharge from 12.2 V to 10 V, otherwise the supply will not properly start. The test consists in either simulating or measuring in the lab how much time the system takes to reach the regulation at full load. Let's suppose that this time corresponds to 6ms. Therefore a VCC fall time of 10 ms could be well appropriated in order to not trigger the overload detection circuitry. If the corresponding IC consumption, including
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NCP1216, NCP1216A
the MOSFET drive, establishes at 2.9 mA, we can calculate the required capacitor using the following formula:
Dt + DV*C i
(eq. 19)
with DV = 2.2 V. Then for a wanted Dt of 30 ms, C equals 39.5 mF or a 68 mF for a standard value (including 20% dispersions). When an overload condition occurs, the IC blocks its internal circuitry and its consumption drops to 350 mA typical. This happens at VCC = 10 V and it remains stuck until VCC reaches 5.6 V: we are in latchoff phase. Again, using the selected 68 mF and 350 mA current consumption, this latchoff phase lasts: 780 ms.
Protecting the Controller Against Negative Spikes
As with any controller built upon a CMOS technology, it is the designer's duty to avoid the presence of negative spikes on sensitive pins. Negative signals have the bad habit to forward bias the controller substrate and induce erratic behaviors. Sometimes, the injection can be so strong that
internal parasitic SCRs are triggered, engendering irremediable damages to the IC if a low impedance path is offered between VCC and GND. If the current sense pin is often the seat of such spurious signals, the high-voltage pin can also be the source of problems in certain circumstances. During the turn-off sequence, e.g. when the user unplugs the power supply, the controller is still fed by its VCC capacitor and keeps activating the MOSFET ON and OFF with a peak current limited by Rsense. Unfortunately, if the quality coefficient Q of the resonating network formed by Lp and Cbulk is low (e.g. the MOSFET Rdson + Rsense are small), conditions are met to make the circuit resonate and thus negatively bias the controller. Since we are talking about ms pulses, the amount of injected charge, (Q = I * t), immediately latches the controller that brutally discharges its VCC capacitor. If this VCC capacitor is of sufficient value, its stored energy damages the controller. Figure 26 depicts a typical negative shot occurring on the HV pin where the brutal VCC discharge testifies for latchup.
0 VCC 5 V/DIV 10 ms/DIV Vlatch 1 V/DIV
Figure 26. A Negative Spike Takes Place on the Bulk Capacitor at the Switch-off Sequence
Simple and inexpensive cures exist to prevent from internal parasitic SCR activation. One of them consists in inserting a resistor in series with the high-voltage pin to keep the negative current to the lowest when the bulk becomes negative (Figure 27). Please note that the negative spike is clamped to (-2 * Vf) due to the diode bridge. Also, the power dissipation of this resistor is extremely small since it only heats up during the startup sequence.
Another option (Figure 28) consists in wiring a diode from VCC to the bulk capacitor to force VCC to reach VCCON sooner and thus stops the switching activity before the bulk capacitor gets deeply discharged. For security reasons, two diodes can be connected in series.
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NCP1216, NCP1216A
Rbulk > 4.7 k + Cbulk 1 2 3 4 8 7 6 5 + CVCC
+ Cbulk 1 2 3 4 8 7 6 5 + CVCC D3 1N4007
Figure 27.
A simple resistor in series avoids any latchup in the controller or one diode forces VCC to reach VCCON sooner.
Figure 28.
Soft-Start - NCP1216A only The NCP1216A features an internal 1.0 ms soft-start activated during the power on sequence (PON). As soon as VCC reaches VCCOFF, the peak current is gradually increased from nearly zero up to the maximum clamping level (e.g. 1.0 V). This situation lasts during 1ms and further to that time period, the peak current limit is blocked to 1.0 V until the supply enters regulation. The soft-start is also
activated during the over current burst (OCP) sequence. Every restart attempt is followed by a soft-start activation. Generally speaking, the soft-start will be activated when VCC ramps up either from zero (fresh power-on sequence) or 5.6 V, the latchoff voltage occurring during OCP. Figure 29 portrays the soft-start behavior. The time scales are purposely shifted to offer a better zoom portion.
Figure 29. Soft-start is activated during a startup sequence or an OCP condition
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NCP1216, NCP1216A
ORDERING INFORMATION
Device NCP1216D65 NCP1216D65G NCP1216D100 NCP1216D133 NCP1216P65 NCP1216P100 NCP1216P133 NCP1216AD65 NCP1216AD100 NCP1216AD133 NCP1216AP65 NCP1216AP100 NCP1216AP133 Version 65 kHz 65 kHz 100 kHz 133 kHz 65 kHz 100 kHz 133 kHz 65 kHz 100 kHz 133 kHz 65 kHz 100 kHz 133 kHz Marking 16D06 16D06 16D10 16D13 P1216P065 P1216P100 P1216P133 16A06 16A10 16A13 P1216AP06 P1216AP10 P1216AP13 Package SOIC-8 SOIC-8 (Pb-Free) SOIC-8 SOIC-8 PDIP-7 PDIP-7 PDIP-7 SOIC-8 SOIC-8 SOIC-8 PDIP-7 PDIP-7 PDIP-7 Shipping 2500 / Tape & Reel 2500 / Tape & Reel 2500 / Tape & Reel 2500 / Tape & Reel 50 Units/ Rail 50 Units/ Rail 50 Units/ Rail 2500 / Tape & Reel 2500 / Tape & Reel 2500 / Tape & Reel 50 Units/ Rail 50 Units/ Rail 50 Units/ Rail
For information on tape and reel specifications, including part orientation and tape sizes, please refer to our Tape and Reel Packaging Specifications Brochure, BRD8011/D.
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NCP1216, NCP1216A
PACKAGE DIMENSIONS
SOIC-8 CASE 751-07 ISSUE AC
-X- A
8 5
B
1 4
S
0.25 (0.010)
M
Y
M
-Y- G C -Z- H D 0.25 (0.010)
M SEATING PLANE
K
NOTES: 1. DIMENSIONING AND TOLERANCING PER ANSI Y14.5M, 1982. 2. CONTROLLING DIMENSION: MILLIMETER. 3. DIMENSION A AND B DO NOT INCLUDE MOLD PROTRUSION. 4. MAXIMUM MOLD PROTRUSION 0.15 (0.006) PER SIDE. 5. DIMENSION D DOES NOT INCLUDE DAMBAR PROTRUSION. ALLOWABLE DAMBAR PROTRUSION SHALL BE 0.127 (0.005) TOTAL IN EXCESS OF THE D DIMENSION AT MAXIMUM MATERIAL CONDITION. 6. 751-01 THRU 751-06 ARE OBSOLETE. NEW STANDARD IS 751-07. MILLIMETERS MIN MAX 4.80 5.00 3.80 4.00 1.35 1.75 0.33 0.51 1.27 BSC 0.10 0.25 0.19 0.25 0.40 1.27 0_ 8_ 0.25 0.50 5.80 6.20 INCHES MIN MAX 0.189 0.197 0.150 0.157 0.053 0.069 0.013 0.020 0.050 BSC 0.004 0.010 0.007 0.010 0.016 0.050 0_ 8_ 0.010 0.020 0.228 0.244
N
X 45 _
0.10 (0.004)
M
J
ZY
S
X
S
DIM A B C D G H J K M N S
SOLDERING FOOTPRINT*
1.52 0.060 7.0 0.275 4.0 0.155
0.6 0.024
1.270 0.050
SCALE 6:1 mm inches
*For additional information on our Pb-Free strategy and soldering details, please download the ON Semiconductor Soldering and Mounting Techniques Reference Manual, SOLDERRM/D.
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NCP1216, NCP1216A
PACKAGE DIMENSIONS
PDIP-7 P SUFFIX CASE 626B-01 ISSUE A
J
8 5
M B L
1
4
NOTES: 1. DIMENSIONS AND TOLERANCING PER ASME Y14.5M, 1994. 2. DIMENSIONS IN MILLIMETERS. 3. DIMENSION L TO CENTER OF LEAD WHEN FORMED PARALLEL. 4. PACKAGE CONTOUR OPTIONAL (ROUND OR SQUARE CORNERS). 5. DIMENSIONS A AND B ARE DATUMS. DIM A B C D F G H J K L M N MILLIMETERS MIN MAX 9.40 10.16 6.10 6.60 3.94 4.45 0.38 0.51 1.02 1.78 2.54 BSC 0.76 1.27 0.20 0.30 2.92 3.43 7.62 BSC --- 10 0.76 1.01
F
NOTE 2
A
C -T-
SEATING PLANE
N D G K B
M
H 0.13 (0.005)
M
TA
M
The product described herein (NCP1216), may be covered by the following U.S. patents: 6,271,735, 6,362,067, 6,385,060, 6,429,709, 6,587,357. There may be other patents pending.
ON Semiconductor and are registered trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC reserves the right to make changes without further notice to any products herein. SCILLC makes no warranty, representation or guarantee regarding the suitability of its products for any particular purpose, nor does SCILLC assume any liability arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including without limitation special, consequential or incidental damages. "Typical" parameters which may be provided in SCILLC data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. All operating parameters, including "Typicals" must be validated for each customer application by customer's technical experts. SCILLC does not convey any license under its patent rights nor the rights of others. SCILLC products are not designed, intended, or authorized for use as components in systems intended for surgical implant into the body, or other applications intended to support or sustain life, or for any other application in which the failure of the SCILLC product could create a situation where personal injury or death may occur. Should Buyer purchase or use SCILLC products for any such unintended or unauthorized application, Buyer shall indemnify and hold SCILLC and its officers, employees, subsidiaries, affiliates, and distributors harmless against all claims, costs, damages, and expenses, and reasonable attorney fees arising out of, directly or indirectly, any claim of personal injury or death associated with such unintended or unauthorized use, even if such claim alleges that SCILLC was negligent regarding the design or manufacture of the part. SCILLC is an Equal Opportunity/Affirmative Action Employer. This literature is subject to all applicable copyright laws and is not for resale in any manner.
PUBLICATION ORDERING INFORMATION
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NCP1216/D

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